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TitleLED Application Design Guide Using Half-Bridge LLC Resonant Converter for 160W Street Lighting
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© 2011 Fairchild Semiconductor Corporation www.fairchildsemi.com
Rev. 1.0.1 • 11/16/12

AN-9730
LED Application Design Guide Using Half-Bridge LLC
Resonant Converter for 160W Street Lighting


Introduction
This application note describes the LED driving system
using a half-bridge LLC resonant converter for high
power LED lighting applications, such as outdoor or
street lighting. Due to the existence of the non-isolation
DC-DC converter to control the LED current and the light
intensity, the conventional PWM DC-DC converter has
the problem of low-power conversion efficiency. The
half-bridge LLC converter can perform the LED current
control and the efficiency can be significantly improved.
Moreover, the cost and the volume of the whole LED
driving system can be reduced.

Consideration of LED Drive
LED lighting is rapidly replacing conventional lighting
sources like incandescent bulbs, fluorescent tubes, and
halogens because LED lighting reduces energy
consumption. LED lighting has greater longevity,
contains no toxic materials, and emits no harmful UV
rays, which are 5 ~ 20 times longer than fluorescent tubes
and incandescent bulbs. All metal halide and fluorescent
lamps, including CFLs, n contain mercury.

The amount of current through an LED determines the
light it emits. The LED characteristics determine the
forward voltage necessary to achieve the required level of
current. Due to the variation in LED voltage versus
current characteristics, controlling only the voltage across
the LED leads to variability in light output. Therefore,
most LED drivers use current regulation to support
brightness control. Brightness can be controlled directly
by changing the LED current.

Consideration of LLC Resonant
Converter
The attempt to obtain ever-increasing power density of
switched-mode power supplies has been limited by the
size of passive components. Operation at higher
frequencies considerably reduces the size of passive
components, such as transformers and filters; however,
switching losses have been an obstacle to high-frequency
operation. To reduce switching losses and allow high-
frequency operation, resonant switching techniques have
been developed. These techniques process power in a
sinusoidal manner and the switching devices are softly
commutated. Therefore, the switching losses and noise
can be dramatically reduced[1-7].

Among various kinds of resonant converters, the simplest
and most popular is the LC series resonant converter, where
the rectifier-load network is placed in series with the L-C
resonant network, as depicted in Figure 1[2-4]. In this
configuration, the resonant network and the load act as a
voltage divider. By changing the frequency of driving
voltage Vd, the impedance of the resonant network changes.
The input voltage is split between this impedance and the
reflected load. Since it is a voltage divider, the DC gain of a
LC series resonant converter is always <1. At light-load
condition, the impedance of the load is large compared to
the impedance of the resonant network; all the input voltage
is imposed on the load. This makes it difficult to regulate
the output at light load. Theoretically, frequency should be
infinite to regulate the output at no load.



Figure 1. Half-Bridge, LC Series Resonant Converter

To overcome the limitation of series resonant converters,
the LLC resonant converter has been proposed[8-12]. The
LLC resonant converter is a modified LC series resonant
converter implemented by placing a shunt inductor across
the transformer primary winding, as depicted in Figure 2.
When this topology was first presented, it did not receive
much attention due to the counterintuitive concept that
increasing the circulating current in the primary side with
a shunt inductor can be beneficial to circuit operation.
However, it can be very effective in improving efficiency
for high-input voltage applications where the switching
loss is more dominant than the conduction loss.

In most practical designs, this shunt inductor is realized
using the magnetizing inductance of the transformer. The
circuit diagram of LLC resonant converter looks much the
same as the LC series resonant converter: the only
difference is the value of the magnetizing inductor. While
the series resonant converter has a magnetizing
inductance larger than the LC series resonant inductor
(Lr), the magnetizing inductance in an LLC resonant
converter is just 3~8 times Lr, which is usually
implemented by introducing an air gap in the transformer.

Page 9

AN-9730 APPLICATION NOTE


© 2011 Fairchild Semiconductor Corporation www.fairchildsemi.com
Rev. 1.0.1 • 11/16/12 8



F
L

S
-X

S


S
e

ri
e

s



Figure 17. Reference Circuit for Design Example of LLC Resonant Half-Bridge Converter



Design Procedure
In this section, a design procedure is presented using the
schematic in Figure 17 as a reference. An integrated
transformer with center tap, secondary side is used and
input is supplied from Power Factor Correction (PFC) pre-
regulator. A DC-DC converter with 160W/115V output
has been selected as a design example. The design
specifications are as follows:

 Nominal input voltage: 400VDC (output of PFC
stage)

 Output: 115V/1.4A (160W)
 Hold-up time requirement: 30ms (50Hz line freq.)
 DC link capacitor of PFC output: 240µF

[STEP-1] Define System Specifications

Estimated Efficiency (Eff): The power conversion
efficiency must be estimated to calculate the maximum
input power with a given maximum output power. If no
reference data is available, use Eff = 0.88~0.92 for low-
voltage output applications and Eff = 0.92~0.96 for high-
voltage output applications. With the estimated efficiency,
the maximum input power is given as:

o

in

ff

P
P

E
 (11)

Input Voltage Range (Vin
min and Vin

max): The maximum
input voltage would be the nominal PFC output voltage as:

max
.in O PFCV V (12)

Even though the input voltage is regulated as constant by
PFC pre-regulator, it drops during the hold-up time. The
minimum input voltage considering the hold-up time
requirement is given as:

min 2
.

2 in HU
in O PFC

DL

P T
V V

C
  (13)

where VO.PFC is the nominal PFC output voltage, THU is
a hold-up time, and CDL is the DC link bulk capacitor.

(Design Example) Assuming the efficiency is 92%,

W
E

P
P

ff

o
in 175

92.0

161


max
. 400in O PFCV V V 

DL

HUin
PFCOin

C

TP
VV

22
.

min 

V341
10240

10301752
400

6

3
2 











[STEP-2] Determine Maximum and Minimum
Voltage Gains of the Resonant Network

As discussed in the previous section, it is typical to operate
the LLC resonant converter around the resonant frequency
(fo) to minimize switching frequency variation. Since the
input of the LLC resonant converter is supplied from PFC
output voltage, the converter should be designed to operate
at fo for the nominal PFC output voltage.

Page 10

AN-9730 APPLICATION NOTE


© 2011 Fairchild Semiconductor Corporation www.fairchildsemi.com
Rev. 1.0.1 • 11/16/12 9

As observed in Equation (10), the gain at fo is a function of
m (m=Lp/Lr). The gain at fo is determined by choosing that
value of m. While a higher peak gain can be obtained with
a small m value, too small m value results in poor coupling
of the transformer and deteriorates the efficiency. It is
typical to set m to be 3~7, which results in a voltage gain
of 1.1~1.2 at the resonant frequency (fo).

With the chosen m value, the voltage gain for the nominal
PFC output voltage is obtained as:

min

1

m
M

m



@f=fo (14)

which would be the minimum gain because the nominal
PFC output voltage is the maximum input voltage (Vin

max).

The maximum voltage gain is given as:

max
max min

min
in

in

V
M M

V
 (15)



(Design Example) The ratio (m) between Lp and Lr is
chosen as 5. The minimum and maximum gains are
obtained as:

12.1
15

5

1
2

max
min 







m

m

V

V
M

in

RO

31.112.1
341

400min
min

max
max  m

V

V
M

in

in

fo

1.12
1

m
M

m
 



fs

Gain (M)

Mmin

Mmax for VIN
min

for
VIN

max

1.31

1.12

Peak Gain
(Available Maximum Gain)

( VO.PFC )


Figure 18. Maximum Gain / Minimum Gain

[STEP-3] Determine the Transformer Turns
Ratio (n=Np/Ns)

With the minimum gain (Mmin) obtained in STEP-2, the
transformer turns ratio is given as:

max
min

2( )
p in

s o F

N V
n M

N V V
  


(16)

where VF is the secondary-side rectifier diode voltage drop.

(Design Example) assuming VF is 0.9V,

93.112.1
)9.0115(2

400

)(2
min

max







 M
VV

V

N

N
n

FO

in

s

p


[STEP-4] Calculate Equivalent Load
Resistance

With the transformer turns ratio obtained from Equation
(16), the equivalent load resistance is obtained as:

2 2

2

8 o
ac

o

n V
R

P
 (17)

(Design Example)








 252

161

9.11593.18)(8
2

222

2

2

 o
Fo

ac
P

VVn
R

[STEP-5] Design the Resonant Network

With m value chosen in STEP-2, read proper Q value from
the peak gain curves in Figure 14 that allows enough peak
gain. Considering the load transient and stable zero-
voltage-switching (ZVS) operation, 10~20% margin
should be introduced on the maximum gain when
determining the peak gain. Once the Q value is
determined, the resonant parameters are obtained as:

1

2
r

o ac

C
Q f R


 

(18)

2

1

(2 )
r

o r

L
f C

 (19)

p rL m L  (20)


(Design Example)
As calculated in STEP-2, the maximum voltage gain
(M max) for the minimum input voltage (Vin

min) is 1.31. With
15% margin, a peak gain of 1.51 is required. m has been
chosen as 5 in STEP-2 and Q is obtained as 0.38 from the
peak gain curves in Figure 19. By selecting the resonant
frequency as 100kHz, the resonant components are
determined as:

nF
acRofQ

rC 64.16
25231010038.02

1

2

1











uH
Cf

L
ro

r 152
1064.16)101002(

1

)2(

1
9232






uHLmL rp 760


Figure 19. Resonant Network Design Using the Peak Gain

(Attainable Maximum Gain)
Curve for m=5

Page 18

AN-9730 APPLICATION NOTE


© 2011 Fairchild Semiconductor Corporation www.fairchildsemi.com
Rev. 1.0.1 • 11/16/12 17

References
[1] Robert L. Steigerwald, “A Comparison of Half-bridge

resonant converter topologies,” IEEE Transactions on
Power Electronics, Vol. 3, No. 2, April 1988.

[2] A. F. Witulski and R. W. Erickson, “Design of the series
resonant converter for minimum stress,” IEEE Transactions
on Aerosp. Electron. Syst., Vol. AES-22, pp. 356-363,
July 1986.

[3] R. Oruganti, J. Yang, and F.C. Lee, “Implementation of
Optimal Trajectory Control of Series Resonant Converters,”
Proc. IEEE PESC ’87, 1987.

[4] V. Vorperian and S. Cuk, “A Complete DC Analysis of the
Series Resonant Converter,” Proc. IEEE PESC’82, 1982.

[5] Y. G. Kang, A. K. Upadhyay, D. L. Stephens, “Analysis and
design of a half-bridge parallel resonant converter operating
above resonance,” IEEE Transactions on Industry
Applications, Vol. 27, March-April 1991, pp. 386 – 395.

[6] R. Oruganti, J. Yang, and F.C. Lee, “State Plane Analysis of
Parallel Resonant Converters,” Proc. IEEE PESC ’85, 1985.

[7] M. Emsermann, “An Approximate Steady State and Small
Signal Analysis of the Parallel Resonant Converter Running
Above Resonance,” Proc. Power Electronics and Variable
Speed Drives ’91, 1991, pp. 9-14.

[8] Yan Liang, Wenduo Liu, Bing Lu, van Wyk, J.D, “Design of
integrated passive component for a 1MHz 1kW half-bridge
LLC resonant converter,” IAS 2005, pp. 2223-2228.

[9] B. Yang, F.C. Lee, M. Concannon, “Over-current protection
methods for LLC resonant converter” APEC 2003, pp. 605 - 609.

[10] Yilei Gu, Zhengyu Lu, Lijun Hang, Zhaoming Qian,
Guisong Huang, “Three-level LLC series resonant DC/DC
converter,” IEEE Transactions on Power Electronics
Vol.20, July 2005, pp.781 – 789.

[11] Bo Yang, Lee, F.C, A.J Zhang, Guisong Huang, “LLC
resonant converter for front-end DC/DC conversion,” APEC
2002. pp.1108 – 1112.

[12] Bing Lu, Wenduo Liu, Yan Liang, Fred C. Lee, Jacobus D.
Van Wyk, “Optimal design methodology for LLC Resonant
Converter,” APEC, 2006, pp.533-538.










This application note written based on Fairchild Semiconductor Application Note AN-4137.






Related Datasheets
FLS2100XS — Half-Bridge LLC Resonant Control IC for Lighting

FAN7346 — 4-Channel LED Current Balance Control IC













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